This invention relates to circuits for reducing distortion produced by an r.f. power amplifier, particularly but not exclusively for SSB (Single Side Band) transmitters.
When an r.f. signal is amplified, any non-linearity in the amplifier will cause envelope amplitude and/or phase distortion of the output compared to the r.f. signal source (drive waveform). This will result in intermodulation distortion of the amplified r.f. signal.
It has been proposed (GB-A-1246209) to compensate for any distortion produced by an r.f. amplifier by separately correcting for envelope amplitude deviations and r.f. phase deviations between the power amplifier output and the drive waveform input.
Referring to FIG. 1a, the inventor attempted to correct for dynamic phase errors between the drive waveform and the output of the power amplifier of a transmitter by means of a phase detector creating an error signal which drives a pre-correcting phase modulator. Any d.c. component of the phase error at the phase detector e.g. that produced even if the power amplifier produced no a.c. phase deviations at all, e.g. due to phase shifts along the cable connecting input B of the phase detector to the probe pick-up P, is not compensated for due to the capacitor C, and this does not matter in many circumstances. The error signal at the output of the difference amplifier performs excursions on the characteristic of the phase detector about an unknown d.c. component of A, B phase difference, a typical characteristic being shown in FIG. 1b, for example, the excursions indicated by X.
However, I have discovered two disadvantages with such an arrangement. First, it would only work if the operating point of the phase detector was on the correct (negative in this example) slope portion of the characteristic. Since the static phase error was unknown, the operating point could be on the wrong (positive in this example) slope portion, and this would result in positive feedback. More importantly, problems were caused with r.f. signals which exhibit cross-overs where there are momentary carrier breaks, e.g. SSB (Single Side Band) transmissions where a two-tone equal amplitude test signal is used. At the carrier breaks, one or both of the inputs A, B of the phase detector would fall to zero, and the phase detector would jump from its arbitrary d.c. level at X to zero, giving a large transient.
To overcome these disadvantages, I introduced a phase shifter and controller (shown dotted) and omitted capacitor C to attempt to hold the A, B phase difference at 90.degree. (as in the phase quadrature detector of GB-A-1246209) i.e. at the operating region Y, so that carrier breaks would not result in a change of output of the phase detector. However, the phase shifter needed to execute phase shifts of up to 360.degree. and operated slowly and in discrete steps. The result was that the phase detector output was held near the origin of FIG. 1b but, depending on the setting of a gain control in the loop between the differential amplifier and the phase modulator, deviated from it by a variable amount of dynamic phase error in operating region Y. However, since a linear multiplier was used for the phase detector (a digital phase detector being impracticable because of the carrier breaks), it turned out that the phase detector output was now dependent on the amplitudes of the signals at A and B as well as their phase. An attempt to incorporate limiters to eliminate the amplitude effects failed because indeterminate signals at the carrier breaks caused large transients. Similarly, the phase difference error signal voltage of GB-A-1246209 is only independent of the amplitudes of the inputs to the phase quadrature detector when those inputs are in quadrature relation, and this will only be true when the loop gain is large. However, the loop gain may not always be large, especially at higher modulation frequencies.